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110 msps/140 msps analog interface for flat panel displays ad9985 rev. 0 information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to chan ge without notice. no license is granted by implication or otherwise under any patent or patent ri ghts of analog devices. trademarks and registered trademarks are the prop erty of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.326.8703 ? 2004 analog devices, inc. all rights reserved. features automated clamping level adjustment 140 msps maximum conversion rate 300 mhz analog bandwidth 0.5 v to 1.0 v analog input range 500 ps p-p pll clock jitter at 110 msps 3.3 v power supply full sync processing sync detect for hot plugging midscale clamping power-down mode low power: 500 mw typical 4:2:2 output format mode applications rgb graphics processing lcd monitors and projectors plasma display panels scan converters microdisplays digital tv functional block diagram r ain r outa g ain g outa b ain b outa midscv sync processing and clock generation dtack hsout vsout sogout ref ref bypass serial register and power management scl sda a0 sogin filt c lamp coas t hsync ad9985 clamp 8 a/d clamp 8 a/d clamp 8 a/d 04799-0-001 auto clamp level adjust auto clamp level adjust auto clamp level adjust figure 1. general description the ad9985 is a complete 8-bit, 140 msps, monolithic analog interface optimized for capturing rgb graphics signals from personal computers and workstations. its 140 msps encode rate capability and full power analog bandwidth of 300 mhz support resolutions up to sxga (1280 1024 at 75 hz). the ad9985 includes a 140 mhz triple adc with internal 1.25 v reference, a pll, and programmable gain, offset, and clamp control. the user provides only a 3.3 v power supply, analog input, and hsync and coast signals. three-state cmos outputs may be powered from 2.5 v to 3.3 v. the ad9985s on-chip pll generates a pixel clock from the hsync input. pixel clock output frequencies range from 12 mhz to 140 mhz. pll clock jitter is 500 ps p-p typical at 140 msps. when the coast signal is presented, the pll maintains its output frequency in the absence of hsync. a sampling phase adjustment is provided. data, hsync, and clock output phase relationships are maintained. the ad9985 also offers full sync processing for composite sync and sync-on-green applications. a clamp signal is generated internally or may be provided by the user through the clamp input pin. this interface is fully programmable via a 2-wire serial interface. fabricated in an advanced cmos process, the ad9985 is provided in a space-saving 80-lead lqfp surface-mount plastic package and is specified over the C40c to +85c temperature range.
ad9985 rev. 0 | page 2 of 32 table of contents revision history ........................................................................... 2 specifications..................................................................................... 3 explanation of test levels........................................................... 6 absolute maximum ratings............................................................ 7 esd caution.................................................................................. 7 pin configuration and function descriptions............................. 8 design guide................................................................................... 11 general description................................................................... 11 digital inputs .............................................................................. 11 input signal handling................................................................ 11 hsync, vsync inputs................................................................... 11 serial control port ..................................................................... 11 output signal handling............................................................. 11 clamping ..................................................................................... 11 rgb clamping........................................................................ 11 yuv clamping ....................................................................... 12 gain and offset control............................................................ 12 auto offset .............................................................................. 12 sync-on-green............................................................................ 13 clock generation ....................................................................... 13 power management.................................................................... 14 timing.......................................................................................... 15 hsync timing ............................................................................. 15 coast timing............................................................................... 15 2-wire serial register map ....................................................... 16 2-wire serial control register detail chip identification... 19 pll divider control .................................................................. 19 clock generator control .......................................................... 19 clamp timing............................................................................. 20 hsync pulsewidth....................................................................... 20 input gain ................................................................................... 20 input offset ................................................................................. 20 mode control 1 .......................................................................... 21 2-wire serial control port........................................................ 26 data transfer via serial interface............................................. 26 sync slicer.................................................................................... 28 sync separator ............................................................................ 28 pcb layout recommendations ............................................... 29 analog interface inputs ............................................................. 29 power supply bypassing ............................................................ 29 pll ............................................................................................... 30 outputs (both data and clocks).............................................. 30 digital inputs .............................................................................. 30 voltage reference ....................................................................... 30 outline dimensions ....................................................................... 31 ordering guide .......................................................................... 31 revision history 5/04revision 0: initial version
ad9985 rev. 0 | page 3 of 32 specifications analog interface: v d = 3.3 v, v dd = 3.3 v, adc clock = maximum conversion rate, unless otherwise noted. table 1. ad9985kstz-110 ad9985kstz-140 parameter temp test level min typ max min typ max unit resolution 8 8 bits dc accuracy differential nonlinearity 25c i 0.5 +1.25/C1.0 0.5 +1.35/?1.0 lsb full vi +1.35/C1.0 1.45/?1.0 lsb integral nonlinearity 25c i 0.5 1.85 0.5 2.0 lsb full vi 2.0 2.3 lsb no missing codes full vi guaranteed guaranteed analog input input voltage range minimum full vi 0.5 0.5 v p-p maximum full vi 1.0 1.0 v p-p gain tempco 25c v 100 100 ppm/c input bias current 25c iv 1 1 a full iv 1 1 a input offset voltage full v 7 7 mv input full-scale matching full vi 1.5 8.0 1.5 8.0 % fs offset adjustment range full vi 46 49 52 46 49 52 % fs reference output output voltage full v 1.25 1.25 v temperature coefficient full v 50 50 ppm/c switching performance maximum conversion rate full vi 110 140 msps minimum conversion rate full iv 10 10 msps data to clock skew full iv ?0.5 +2.0 ?0.5 +2.0 ns t buff full vi 4.7 4.7 s t stah full vi 4.0 4.0 s t dho full vi 300 300 ns t dal full vi 4.7 4.7 s t dah full vi 4.0 4.0 s t dsu full vi 250 250 ns t stasu full vi 4.7 4.7 s t stotsu full vi 4.0 4.0 s hsync input frequency full iv 15 110 15 110 khz maximum pll clock rate full vi 110 140 mhz minimum pll clock rate full iv 12 12 mhz pll jitter 25c iv 400 700 1 400 700 1 ps p-p full iv 1000 1 400 700 1 ps p-p sampling phase tempco full iv 15 15 ps/c digital inputs input voltage, high (v ih ) full vi 2.5 2.5 v input voltage, low (v il ) full vi 0.8 0.8 v input current, high (v ih ) full v ?1.0 ?1.0 a input current, low (v il ) full v +1.0 +1.0 a input capacitance 25c v 3 3 pf
ad9985 rev. 0 | page 4 of 32 ad9985kstz-110 ad9985kstz-140 parameter temp test level min typ max min typ max unit digital outputs output voltage, high (v oh ) full vi v d ?0.1 v d ?0.1 v output voltage, low (v ol ) full vi 0.1 0.1 v duty cycle datack full iv 45 50 55 45 50 55 % output coding binary binary power supply v d supply voltage full iv 3.15 3.3 3.45 3.15 3.3 3.45 v v dd supply voltage full iv 2.2 3.3 3.45 2.2 3.3 3.45 v p vd supply voltage full iv 3.15 3.3 3.45 3.15 3.3 3.45 v i d supply current (v d ) 25c v 132 180 ma i dd supply current (v dd ) 2 25c v 19 26 ma ip vd supply current (p vd ) 25c v 8 11 ma total power dissipation full vi 525 760 650 900 mw power-down supply current full vi 5 15 5 15 ma power-down dissipation full vi 16.5 50 16.5 50 mw dynamic performance analog bandwidth, full power 25c v 300 300 mhz transient response 25c v 2 2 ns overvoltage recovery time 25c v 1.5 1.5 ns signal-to-noise ratio (snr) 25c v 44 43 db (without harmonics) full v 43 42 db f in = 40.7 mhz crosstalk full v 55 55 dbc thermal characteristics jc junction-to-case thermal resistance v 16 16 c/w ja junction-to-ambient thermal resistance v 35 35 c/w 1 vco range = 10, charge pump current = 110, pll divider = 1693. 2 datack load = 15 pf, data load = 5 pf.
ad9985 rev. 0 | page 5 of 32 table 2. ad9985bstz-110 parameter temp test level min typ max unit resolution 8 bits dc accuracy lsb differential nonlinearity 25c i 0.5 +1.25/?1.0 lsb full vi +1.5/?1.0 lsb integral nonlinearity 25c i 0.5 1.85 lsb full vi 3.2 analog input input voltage range minimum full vi 0.5 v p-p maximum full vi 1.0 v p-p gain tempco 25c v 100 ppm/c input bias current 25c iv 1 a full iv 2 a input offset voltage full vi 7 mv input full-scale matching full vi 1.5 8.0 % fs offset adjustment range full vi 46 49 52 % fs reference output output voltage full vi 1.25 v temperature coefficient full v 100 ppm/c switching performance maximum conversion rate full vi 110 msps minimum conversion rate full iv 10 msps data to clock skew full iv C0.5 +2.0 ns t buff full vi 4.7 s t stah full vi 4.0 s t dho full vi 300 ns t dal full vi 4.7 s t dah full vi 4.0 s t dsu full vi 250 ns t stasu full vi 4.7 s t stah full vi s hsync input frequency full iv 15 110 khz maximum pll clock rate full vi 110 mhz minimum pll clock rate full iv 12 mhz pll jitter 25c iv 400 700 1 ps p-p full iv 1000 1 ps p-p sampling phase tempco full iv 15 ps/c digital inputs input voltage, high (v ih ) full vi 2.5 v input voltage, low (v il ) full vi 0.8 v input current, high (i ih ) full v ?1.0 a input current, low (i il ) full v 1.0 a input capacitance 25c v 3 pf digital outputs output voltage, high (v oh ) full vi v d ?0.1 v output voltage, low (v ol ) full vi 0.1 v duty cycle, datack full iv 45 50 55 % output coding binary
ad9985 rev. 0 | page 6 of 32 ad9985bstz-110 parameter temp test level min typ max unit power supply v d supply voltage full iv 3.15 3.3 3.45 v v dd supply voltage full iv 2.2 3.3 3.45 v p vd supply voltage full iv 3.15 3.3 3.45 v i d supply current (v d ) 25c v 132 ma i dd supply current (v dd ) 2 25c v 19 ma ip vd supply current (p vd ) 25c v 8 ma total power dissipation full vi 525 760 mw power-down supply current full vi 5 15 ma power-down dissipation full vi 16.5 50 mw dynamic performance analog bandwidth, full power 25c v 300 mhz transient response 25c v 2 ns overvoltage recovery time 25c v 1.5 ns signal-to-noise ratio (snr) 25c v 44 db (without harmonics) full v 43 db f in = 40.7 mhz crosstalk full v 55 dbc thermal characteristics jc junction-to-case thermal resistance v 16 c/w ja junction-to-ambient thermal resistance v 35 c/w 1 vco range = 10, charge pump current = 110, pll divider = 1693. 2 datack load = 15 pf, data load = 5 pf. . explanation of test levels te s t l e ve l i. 100% production tested. ii. 100% production tested at 25c and sample tested at specified temperatures. iii. sample tested only. iv. parameter is guaranteed by design and characterization testing. v. parameter is a typical value only. vi. 100% production tested at 25c; guaranteed by design and characterization testing.
ad9985 rev. 0 | page 7 of 32 absolute maximum ratings table 3. parameter rating v d 3.6 v v dd 3.6 v analog inputs v d to 0.0 v vref in v d to 0.0 v digital inputs 5 v to 0.0 v digital output current 20 ma operating temperature ?40c to +85c storage temperature ?65c to +150c maximum junction temperature 150c maximum case temperature 150c stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions outside of those indicated in the operational sections of this specification is not implied. exposure to absolute maximum ratings for extended periods may affect device reliability. esd caution esd (electrostatic discharge) sensitive device. electrosta tic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge wi thout detection. although this product features proprietary esd protection circuitry, permanent dama ge may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality.
ad9985 rev. 0 | page 8 of 32 pin configuration and fu nction descriptions 04799-0-002 80 79 78 77 76 71 70 69 68 75 74 73 72 21 22 23 24 25 26 27 28 29 30 31 32 33 1 2 3 4 5 6 7 8 9 10 11 13 12 60 59 58 57 56 55 54 53 52 51 50 49 48 ad9985 top view (not to scale) gnd v dd v dd gnd gnd pv d pv d gnd coast hsync vsync gnd filt gnd v dd v dd red <0> red <1> red <2> red <3> red <4> red <5> red <6> red <7> v dd gnd gnd g reen <7> g reen <6> g reen <5> g reen <4> g reen <3> g reen <2> g reen <1> g reen <0> gnd v dd blue <7> blue <6> gnd v d ref bypass sda scl a0 r ain gnd v d v d gnd sogin g ain pin 1 indicator 14 15 16 17 18 20 19 47 46 45 44 43 42 41 blue <5> blue <4> blue <3> blue <2> blue <1> blue <0> gnd gnd v d v d gnd b ain v d gnd 64 63 62 61 67 66 65 34 35 36 37 38 39 40 pv d pv d gnd midscv clamp v d gnd datack hsout sogou t vsout gnd v d gnd figure 2. pin configuration table 4. complete pinout list pin type mnemonic function value pin no. inputs r ain analog input for conver ter r 0.0 v to 1.0v 54 g ain analog input for converter g 0.0 v to 1.0v 48 b ain analog input for converter b 0.0 v to 1.0v 43 hsync horizontal sync input 3.3 v cmos 30 vsync vertical sync input 3.3 v cmos 31 sogin input for sync-on-green 0.0 v to 1.0 v 49 clamp clamp input (external clamp signal) 3.3 v cmos 38 coast pll coast signal input 3.3 v cmos 29 outputs red [7:0] outputs of converter red, bit 7 is the msb 3.3 v cmos 70C77 green [7:0] outputs of converter gr een, bit 7 is the bsb 3.3 v cmos 2C9 blue [7:0] outputs of converter bl ue, bit 7 is the bsb 3.3 v cmos 12C19 datack data output clock 3.3 v cmos 67 hsout hsync output (phase-aligned with datack) 3.3 v cmos 66 vsout vsync output (phase-aligned with datack) 3.3 v cmos 64 sogout sync-on-green slicer output 3.3 v cmos 65 references ref bypass internal reference bypass 1.25 v 58 midscv internal midscale voltage bypass 37 filt connection for external filter components for internal pll 33 power supply v d analog power supply 3.3 v 39, 42, 45, 46, 51, 52, 59, 62 v dd output power supply 3.3 v 11, 22, 23, 69, 78, 79 pv d pll power supply 3.3 v 26, 27, 34, 35 gnd ground 0 v 1, 10, 20, 21, 24, 25, 28, 32, 36, 40, 41, 44, 47, 50, 53, 60, 61, 63, 68, 80 control sda serial port data i/o 3.3 v cmos 57 scl serial port data clock (100 khz maximum 3.3 v cmos 56 a0 serial port address input 1 3.3 v cmos 55
ad9985 rev. 0 | page 9 of 32 table 5. pin function descriptions pin name function outputs hsout horizontal sync output a reconstructed and phase-aligned version of the hsync input. both the polari ty and duration of this output can be programmed via serial bus registers. by maintaining alignment wi th datack and data, data timin g with respect to horizontal sync can always be determined. vsout vertical sync output a reconstructed and phase-aligned version of the video vsync. the polarity of this o utput can be controlled via a serial bus bi t. the placement and duration in all mode s is set by the graphics transmitter. sogout sync-on-green slicer output this pin outputs either the signal from the sync-on-green slicer comparator or an unprocessed but delayed version of the hsync input. see the sync processing block diagram (figure 14) to view how this pin is connected. (note: besides slicing off sog, the output from this pin gets no ot her additional processing on the ad9985. vs ync separation is performed via the sync separator.) serial port (2-wire) sda serial port data i/o scl serial port data clock a0 serial port address input 1 for a full description of the 2-wire serial register and how it works, refer to the 2-wire serial control port section. data outputs red data output, red channel green data output, green channel blue data output, blue channel the main data outputs. bit 7 is the msb. the delay from pixel sampling time to output is fixed. when the sampling time is changed by adjusting the phase register, the output timing is shifted as well. th e datack and hsout outputs are also moved, so the timing relationship among the signals is maintained . for exact timing information, re fer to figure 9, figure 10, and figure 11. data clock output datack data output clock the main clock output signal used to strobe the output data and hsou t into external logic. it is produced by the internal clock generator and is synchronous with the inte rnal pixel sampling clock. when the samp ling time is changed by adjusting the phase register, the output timing is shif ted as well. the data, datack, and hsou t outputs are all moved, so the timing relationship among the signals is maintained. inputs r ain analog input for red channel g ain analog input for green channel b ain analog input for blue channel high impedance inputs that acce pt the red, green, and blue channel graphics signals, respectively. (the three channels are identical, and can be used for any colors , but colors are assigned for convenient reference.) they acco mmodate input signals ranging from 0.5 v to 1.0 v full scale. signals should be ac-coupled to these pins to support clamp operation. hsync horizontal sync input this input receives a logic signal that establishes the horizo ntal timing reference and provides the frequency reference for pi xel clock generation. the logic sense of this pin is controlled by serial register 0eh bit 6 (hsync polarity). only the leading edg e of hsync is active; the trailing edge is igno red. when hsync polarity = 0, the falling ed ge of hsync is used . when hsync polarity = 1, the rising edge is active. the input includes a schmitt trigger for noise immunity, with a nominal input threshold of 1.5 v. vsync vertical sync input the input for vertical sync.
ad9985 rev. 0 | page 10 of 32 pin name function sogin sync-on-green input this input is provided to assist with processing signals wi th embedded sync, typically on the green channel. the pin is connected to a high speed comparator with an internally ge nerated threshold. the threshold level can be programmed in 10 mv steps to any voltage between 10 mv and 330 mv above th e negative peak of the input signal. the default voltage threshold is 150 mv. when connected to an ac-coupled graphics signal with embe dded sync, it will produce a noninverting digital output on sogout. (this is usually a composite sync signal, containing both vertical and horizontal sync information that must be separated before passing the horizontal sync si gnal to hsync.) when not used , this input should be left unconnected. for more details on this fu nction and how it should be configured, refer to the sync-on-green section. clamp external clamp input this logic input may be used to define the time during which the input signal is clamped to ground. it should be exercised when the reference dc level is known to be present on the an alog input channels, typically during the back porch of the graphics signal. the clamp pin is enabled by setting control bit clamp function to 1 (register 0f h, bit 7, default is 0). when disabled, this pin is ignored and the clamp timing is determined internally by counting a delay and duration from the trailing edge of the hsync input. the logic sense of this pin is controll ed by clamp polarity register 0fh, bit 6. when not used, this p in must be grounded and clamp function programmed to 0. coast clock generator coast input (optional) this input may be used to cause the pixel clock generator to stop synchroniz ing with hsync and continue producing a clock at its current frequency and phase. this is usef ul when processing signals from sources that fail to produce horizontal sync pulse s during the vertical interval. th e coast signal is generally not required for pc-generated signals. the logic sense of this pin is controlled by coast polarity (reg ister 0fh, bit 3). when not used, this pin ma y be grounded and coas t polarity programmed to 1, or tied high (to v d through a 10 k? resistor) and coast polarity programme d to 0. coast polarity defa ults to 1 at power-up. ref bypass internal reference bypass bypass for the internal 1.25 v band gap reference. it should be connected to ground through a 0. 1 f capacitor. the absolute accuracy of this reference is 4%, and the temperature coe fficient is 50 ppm, which is ad equate for most ad9985 applica- tions. if higher accuracy is required, an external reference may be employed instead. midscv midscale voltage reference bypass bypass for the internal midscale voltage reference. it should be connected to ground through a 0.1 f capacitor. the exact voltage varies with the gain setting of the blue channel. filt external filter connection for proper operation, the pixel clock generato r pll requires an external filter. connect the filter shown in figure 8 to this p in. for optimal performance, minimize noise and parasitics on this node. power supply v d main power supply these pins supply power to the main elements of the circ uit. they should be filtered and as quiet as possible. v dd digital output power supply a large number of output pins (up to 25) switching at high sp eed (up to 110 mhz) generates a lot of power supply transients (noise). these supply pins are identified separately from the v d pins so special care can be taken to minimize output noise transferred into the sensitive analog circuitry. if the ad9985 is interfacing with lower voltage logic, v dd may be connected to a lower supply voltage (as low as 2.5 v) for compatibility. pv d clock generator power supply the most sensitive portion of the ad9985 is the clock generation circuitry. these pins provide power to the clock pll and help the user design for optimal performance. the designer should provide quiet, noise-free power to these pins. gnd ground the ground return for all circuitry on-chi p. it is recommended that the ad9985 be assembled on a single solid ground plane, with careful attention given to ground current paths.
ad9985 rev. 0 | page 11 of 32 design guide general description the ad9985 is a fully integrated solution for capturing analog rgb signals and digitizing them for display on flat-panel monitors or projectors. the circuit is ideal for providing a computer interface for hdtv monitors or as the front end to high performance video scan converters. implemented in a high performance cmos process, the interface can capture signals with pixel rates up to 110 mhz. the ad9985 includes all necessary input buffering, signal dc restoration (clamping), offset and gain (brightness and contrast) adjustment, pixel clock generation, sampling phase control, and output data formatting. all controls are programmable via a 2-wire serial interface. full integration of these sensitive analog functions makes system design straightforward and less sensitive to the physical and electrical environment. with a typical power dissipation of only 500 mw and an operating temperature range of 0c to 70c, the device requires no special environmental considerations. digital inputs all digital inputs on the ad9985 operate to 3.3 v cmos levels. however, all digital inputs are 5 v tolerant. applying 5 v to them will not cause any damage. input signal handling the ad9985 has three high impedance analog input pins for the red, green, and blue channels. they will accommodate signals ranging from 0.5 v to 1.0 v p-p. signals are typically brought onto the interface board via a dvi-i connector, a 15-pin d connector, or via bnc connectors. the ad9985 should be located as close as practical to the input connector. signals should be routed via matched-impedance traces (normally 75 ?) to the ic input pins. at that point the signal should be resistively terminated (75 ? to the signal ground return) and capacitively coupled to the ad9985 inputs through 47 nf capacitors. these capacitors form part of the dc restoration circuit. in an ideal world of perfectly matched impedances, the best performance can be obtained with the widest possible signal bandwidth. the ultrawide bandw idth inputs of the ad9985 (300 mhz) can track the input signal continuously as it moves from one pixel level to the next, and digitize the pixel during a long, flat pixel time. in many systems, however, there are mismatches, reflections, and noise, which can result in excessive ringing and distortion of the input waveform. this makes it more difficult to establish a sampling phase that provides good image quality. it has been shown that a small inductor in series with the input is effective in rolling off the input bandwidth slightly and providing a high quality signal over a wider range of conditions. using a fair-rite #2508051217z0 high speed signal chip bead inductor in the circuit of figure 3 gives good results in most applications. rgb input r ain g ain b ain 47nf 75 ? 04799-0-003 figure 3. analog input interface circuit hsync, vsync inputs the interface also takes a horizontal sync signal, which is used to generate the pixel clock and clamp timing. this can be either a sync signal directly from the graphics source, or a preproc- essed ttl or cmos level signal. the hsync input includes a schmitt trigger buffer for immunity to noise and signals with long rise times. in typical pc-based graphic systems, the sync signals are simply ttl-level drivers feeding unshielded wires in the monitor cable. as such, no termination is required. serial control port the serial control port is designed for 3.3 v logic. if there are 5 v drivers on the bus, these pins should be protected with 150 ? series resistors placed between the pull-up resistors and the input pins. output signal handling the digital outputs are designed and specified to operate from a 3.3 v power supply (v dd ). they can also work with a v dd as low as 2.5 v for compatibility with other 2.5 v logic. clamping rgb clamping to properly digitize the incoming signal, the dc offset of the input must be adjusted to fit the range of the on-board a/d converters. most graphics systems produce rgb signals with black at ground and white at approximately 0.75 v. however, if sync signals are embedded in the graphics, the sync tip is often at ground and black is at 300 mv. then white is at approximately 1.0 v. some common rgb line amplifier boxes use emitter- follower buffers to split signals and increase drive capability. this introduces a 700 mv dc offset to the signal, which must be removed for proper capture by the ad9985. the key to clamping is to identify a portion (time) of the signal when the graphic system is known to be producing black. an offset is then introduced which results in the a/d converters producing a black output (code 00h) when the known black
ad9985 rev. 0 | page 12 of 32 input is present. the offset then remains in place when other signal levels are processed, and the entire signal is shifted to eliminate offset errors. in most pc graphics systems, black is transmitted between active video lines. with crt displays, when the electron beam has completed writing a horizontal line on the screen (at the right side), the beam is deflected quickly to the left side of the screen (called horizontal retrace), and a black signal is provided to prevent the beam from disturbing the image. in systems with embedded sync, a blacker-than-black signal (hsync) is produced briefly to signal the crt that it is time to begin a retrace. for obvious reasons, it is important to avoid clamping on the tip of hsync. fortunately, there is virtually always a period following hsync, called the back porch, where a good black reference is provided. this is the time when clamping should be done. the clamp timing can be established by simply exercising the clamp pin at the appropriate time (with external clamp = 1). the polarity of this signal is set by the clamp polarity bit. a simpler method of clamp timing employs the ad9985 internal clamp timing generator. the clamp placement register is programmed with the number of pixel times that should pass after the trailing edge of hsync before clamping starts. a second register (clamp duration) sets the duration of the clamp. these are both 8-bit values, providing considerable flexibility in clamp generation. the clamp timing is referenced to the trailing edge of hsync because, though hsync duration can vary widely, the back porch (black reference) always follows hsync. a good starting point for establishing clamping is to set the clamp placement to 09h (providing 9 pixel periods for the graphics signal to stabilize after sync) and set the clamp duration to 14h (giving the clamp 20 pixel periods to reestablish the black reference). clamping is accomplished by placing an appropriate charge on the external input coupling capacitor. the value of this capacitor affects the performance of the clamp. if it is too small, there will be a significant amplitude change during a horizontal line time (between clamping intervals). if the capacitor is too large, then it will take excessively long for the clamp to recover from a large change in incoming signal offset. the recommended value (47 nf) results in recovering from a step error of 100 mv to within 1/2 lsb in 10 lines with a clamp duration of 20 pixel periods on a 60 hz sxga signal. yuv clamping yuv graphic signals are slightly different from rgb signals in that the dc reference level (black level in rgb signals) can be at the midpoint of the graphics signal rather than at the bottom. for these signals, it can be necessary to clamp to the midscale range of the a/d converter range (80h) rather than at the bottom of the a/d converter range (00h). clamping to midscale rather than to ground can be accom- plished by setting the clamp select bits in the serial bus register. each of the three converters has its own selection bit so that they can be clamped to either midscale or ground inde- pendently. these bits are located in register 10h and are bits 0C2. the midscale reference voltage that each a/d converter clamps to is provided on the midscv pin (pin 37). this pin should be bypassed to ground with a 0.1 f capacitor, even if midscale clamping is not required. gain 1.0 0 00h ffh input range (v) 0.5 offset = 00h offset = 3fh offset = 7fh offset = 00h offset = 7fh offset = 3fh 04799-0-004 figure 4. gain and offset control gain and offset control the ad9985 can accommodate input signals with inputs ranging from 0.5 v to 1.0 v full scale. the full-scale range is set in three 8-bit registers (red gain, green gain, and blue gain). note that increasing the gain setting results in an image with less contrast. the offset control shifts the entire input range, resulting in a change in image brightness. three 7-bit registers (red offset, green offset, blue offset) provide independent settings for each channel. the offset controls provide a 63 lsb adjustment range. this range is connected with the full-scale range, so if the input range is doubled (from 0.5 v to 1.0 v) then the offset step size is also doubled (from 2 mv per step to 4 mv per step). figure 4 illustrates the interaction of gain and offset controls. the magnitude of an lsb in offset adjustment is proportional to the full-scale range, so changing the full-scale range also changes the offset. the change is minimal if the offset setting is near midscale. when changing the offset, the full-scale range is not affected, but the full-scale level is shifted by the same amount as the zero-scale level. auto offset in addition to the manual offset adjustment mode (via registers 0bh to 0dh), the ad9985 also includes circuitry to automatically calibrate the offset for each channel. by monitoring the output of each adc during the back porch of the input signals, the ad9985 can self-adjust to eliminate any
ad9985 rev. 0 | page 13 of 32 offset errors in its own adc channels as well as any offset errors present on the incoming graphics or video signals. to activate the auto-offset mode, set register 1dh, bit 7 to 1. next, the target code registers (19h through 1bh) must be programmed. the values programmed into the target code registers should be the output code desired from the ad9985 during the back porch reference time. for example, for rgb signals, all three registers would normally be programmed to code 1, while for ypbpr signals the green (y) channel would normally be programmed to code 1 and the blue and red channels (pb and pr) would normally be set to 128. any target code value between 1 and 254 can be set, although the ad9985s offset range may not be able to reach every value. intended target code values range from (but are not limited to) 1 to 40 when ground clamping and 90 to 170 when midscale clamping. the ability to program a target code for each channel gives users a large degree of freedom and flexibility. while in most cases all channels will be set to either 1 or 128, the flexibility to select other values allows for the possibility of inserting intentional skews between channels. it also allows for the adc range to be skewed so that voltages outside of the normal range can be digitized. (for example, setting the target code to 40 would allow the sync tip, which is normally below black level, to be digitized and evaluated.) lastly, when in auto offset mode, the manual offset registers (0bh to 0dh) have new functionality. the values in these registers are digitally added to the value of the adc output. the purpose of doing this is to match a benefit that is present with manual offset adjustment. adjusting these registers is an easy way to make brightness adjustments. although some signal range is lost with this method, it has proven to be a very popular function. in order to be able to increase and decrease brightness, the values in these registers in this mode are signed twos complement. the digital adder is used only when in auto offset mode. although it cannot be disabled, setting the offset registers to all 0s will effectively disable it by always adding 0. sync-on-green the sync-on-green input operates in two steps. first, it sets a baseline clamp level off of the incoming video signal with a negative peak detector. second, it sets the sync trigger level to a programmable level (typically 150 mv) above the negative peak. the sync-on-green input must be ac-coupled to the green analog input through its own capacitor, as shown in figure 5. the value of the capacitor must be 1 nf 20%. if sync-on- green is not used, this connection is not required. note that the sync-on-green signal is always negative polarity. r ain b ain g ain sog 47nf 47nf 47nf 1nf 04799-0-005 figure 5. typical clamp configuration clock generation a phase-locked loop (pll) is employed to generate the pixel clock. in this pll, the hsync input provides a reference frequency. a voltage controlled oscillator (vco) generates a much higher pixel clock frequency. this pixel clock is divided by the pll divide value (registers 01h and 02h) and phase compared with the hsync input. any error is used to shift the vco frequency and maintain lock between the two signals. the stability of this clock is a very important element in providing the clearest and most stable image. during each pixel time, there is a period during which the signal is slewing from the old pixel amplitude and settling at its new value. then there is a time when the input voltage is stable, before the signal must slew to a new value (figure 6). the ratio of the slewing time to the stable time is a function of the bandwidth of the graphics dac and the bandwidth of the transmission system (cable and termination). it is also a function of the overall pixel rate. clearly, if the dynamic characteristics of the system remain fixed, the slewing and settling time is likewise fixed. this time must be subtracted from the total pixel period, leaving the stable period. at higher pixel frequencies, the total cycle time is shorter, and the stable pixel time becomes shorter as well. pixel clock invalid sample times 04799-0-006 figure 6. pixel sampling times any jitter in the clock reduces the precision with which the sampling time can be determined, and must also be subtracted from the stable pixel time. considerable care has been taken in the design of the ad9985s clock generation circuit to minimize jitter. as indicated in figure 7, the clock jitter of the ad9985 is less than 5% of the total pixel time in all operating modes, making the reduction in the valid sampling time due to jitter negligible.
ad9985 rev. 0 | page 14 of 32 frequency (mhz) 14 12 0 0 pixel clock jitter (p-p) (%) 10 8 6 4 2 10 20 30 40 50 60 70 80 90 100 110 120 130 140 150 04799-0-007 figure 7. pixel clock jitter vs. frequency the pll characteristics are determined by the loop filter design, by the pll charge pump current, and by the vco range setting. the loop filter design is illustrated in figure 8. recommended settings of vco range and charge pump current for vesa standard display modes are listed in table 9. c p 0.0082 f c z 0.082 f r z 2.7k ? filt pv d 04799-0-008 figure 8. pll loop filter detail four programmable registers are provided to optimize the performance of the pll: 1. the 12-bit divisor register. the input hsync frequencies range from 15 khz to 110 khz. the pll multiplies the frequency of the hsync signal, producing pixel clock frequencies in the range of 12 mhz to 110 mhz. the divisor register controls the exact multiplication factor. this register may be set to any value between 221 and 4095. (the divide ratio that is actually used is the programmed divide ratio plus one.) 2. the 2-bit vco range register. to improve the noise performance of the ad9985, the vco operating frequency range is divided into three overlapping regions. the vco range register sets this operating range. table 6 lists the frequency ranges for the lowest and highest regions. table 6. vco frequency ranges pixel clock range (mhz) pv1 pv0 ad9985kstz ad9985bstz 0 0 12C32 12C30 0 1 32C64 30C60 1 0 64C110 60C110 1 1 110C140 3. the 3-bit charge pump current register. this register allows the current that drives the low-pass loop filter to be varied. the possible current values are listed in table 7. table 7. charge pump current/control bits ip2 ip1 ip0 current (a) 0 0 0 50 0 0 1 100 0 1 0 150 0 1 1 250 1 0 0 350 1 0 1 500 1 1 0 750 1 1 1 1500 4. the 5-bit phase adjust register. the phase of the gen- erated sampling clock may be shifted to locate an optimum sampling point within a clock cycle. the phase adjust register provides 32 phase-shift steps of 11.25 each. the hsync signal with an identical phase shift is available through the hsout pin. the coast pin is used to allow the pll to continue to run at the same frequency, in the absence of the incoming hsync signal or during disturbances in hsync (such as equalization pulses). this may be used during the vertical sync period, or any other time that the hsync signal is unavailable. the polarity of the coast signal may be set through the coast polarity register. also, the polarity of the hsync signal may be set through the hsync polarity register. if not using automatic polarity detection, the hsync and coast polarity bits should be set to match the respective polarities of the input signals. power management the ad9985 uses the activity detect circuits, the active interface bits in the serial bus, the active interface override bits, and the power-down bit to determine the correct power state. there are three power statesfull-power, seek mode, and power-down. table 8 summarizes how the ad9985 determines what power mode to be in and which circuitry is powered on/off in each of these modes. the power-down command has priority over the automatic circuitry. table 8. power-down mode descriptions mode inputs power-down 1 sync detect 2 powered on or comments full- power 1 1 everything seek mode 1 0 serial bus, sync activity detect, sog, band gap reference power- down 0 x serial bus, sync activity detect, sog, band gap reference 1 power-down is controlled via bit 1 in serial bus register 0fh. 2 sync detect is determined by oring bits 7, 4, and 1 in serial bus register 14h.
ad9985 rev. 0 | page 15 of 32 table 9. recommended vco range and charge pump current settings for standard display formats ad9985kstz ad9985bstz standard modes resolution refresh rate (hz) horizontal frequency (khz) pixel rate (mhz) pll div vcornge current vcornge current vga 640 480 60 31.5 25.175 799 00 110 00 011 72 37.7 31.500 835 00 110 01 010 75 37.5 31.500 841 00 110 01 010 85 43.3 36.000 831 01 100 01 010 svga 800 600 56 35.1 36.000 1025 01 100 01 010 60 37.9 40.000 1055 01 100 01 011 72 48.1 50.000 1039 01 101 01 100 75 46.9 49.500 1055 01 101 01 100 85 53.7 56.250 1047 01 101 01 101 xga 1024 768 60 48.4 65.000 1343 10 101 10 011 70 56.5 75.000 1327 10 100 10 011 75 60.0 78.750 1313 10 100 10 011 80 64.0 85.500 1335 10 101 10 100 85 68.3 94.500 1383 10 101 10 100 sxga 1280 1024 60 64.0 108.000 1687 10 110 10 101 75 80.0 135.000 1687 11 110 tv modes 480i 720 480 60 15.75 13.51 857 00 011 00 011 480p 720 483 60 31.47 27.00 857 00 110 00 011 720p 1280 720 60 45.0 74.25 1649 10 100 10 011 1080i 1920 1080 60 33.75 74.25 2199 10 100 10 011 timing the following timing diagrams show the operation of the ad9985. the output data clock signal is created so that its rising edge always occurs between data transitions and can be used to latch the output data externally. there is a pipeline in the ad9985, which must be flushed before valid data becomes available. this means that four data sets are presented before valid data is available. t per t cycle t skew datac k data hsout 04799-0-009 figure 9. output timing hsync timing horizontal sync (hsync) is processed in the ad9985 to eliminate ambiguity in the timing of the leading edge with respect to the phase-delayed pixel clock and data. the hsync input is used as a reference to generate the pixel sampling clock. the sampling phase can be adjusted, with respect to hsync, through a full 360 in 32 steps via the phase adjust register (to optimize the pixel sampling time). display systems use hsync to align memory and display write cycles, so it is important to have a stable timing relationship between hsync output (hsout) and data clock (datack). three things happen to horizontal sync in the ad9985. first, the polarity of hsync input is determined and will thus have a known output polarity. the known output polarity can be programmed either active high or active low (register 0eh, bit 5). second, hsout is aligned with datack and data outputs. third, the duration of hsout (in pixel clocks) is set via register 07h. hsout is the sync signal that should be used to drive the rest of the display system. coast timing in most computer systems, the hsync signal is provided continuously on a dedicated wire. in these systems, the coast input and function are unnecessary and should not be used, and the pin should be permanently connected to the inactive state. in some systems, however, hsync is disturbed during the vertical sync period (vsync). in some cases, hsync pulses
ad9985 rev. 0 | page 16 of 32 disappear. in other systems, such as those that employ composite sync (csync) signals or embedded sync-on-green (sog), hsync includes equalization pulses or other distortions during vsync. to avoid upsetting the clock generator during vsync, it is important to ignore these distortions. if the pixel clock pll sees extraneous pulses, it will attempt to lock to this new frequency, and will have changed frequency by the end of the vsync period. it will then take a few lines of correct hsync timing to recover at the beginning of a new frame, resulting in a tearing of the image at the top of the display. the coast input is provided to eliminate this problem. it is an asynchronous input that disables the pll input and allows the clock to free-run at its then-current frequency. the pll can free-run for several lines without significant frequency drift. . p0 p1 p2 p3 p4 p5 p6 p7 5-pipe delay d0 d1 d2 d3 d4 d5 d6 d7 rgb in hsync pxck hs adcck datack d outa hsout variable duration 04799-0-010 figure 10. 4:4:4 mode (for rgb and yuv) p0 p1 p2 p3 p4 p5 p6 p7 5-pipe delay y0 y1 y2 y3 y4 y5 y6 y7 rgb in hsync pxck hs adcck datac k g outa hsout u0 v1 u2 v3 u4 v5 u6 v7 r outa variable duration 04799-0-011 figure 11. 4:2:2 mode (for yuv only) 2-wire serial register map the ad9985 is initialized and controlled by a set of registers, th at determine the operating modes. an external controller is e mployed to write and read the control registers through the two-line serial interface port. table 10. control register map hex address write and read or read only bits default value register name function 00h ro 7:0 chip revision an 8-bit register that represents the silicon revision level. 01h* r/w 7:0 01101001 pll div msb this register is for bits [ 11:4] of the pll divider. greater values mean the pll operates at a faster rate. this register should be loaded first whenever a change is needed. this wi ll give the pll more time to lock. 02h* r/w 7:4 1101**** pll div lsb bits [7:4] of this word are written to the lsbs [3:0] of the pll divider word.
ad9985 rev. 0 | page 17 of 32 hex address write and read or read only bits default value register name function 03h r/w 7:3 01****** bits [7:6] vco range. selects vco frequency range. (see pll description.) **001*** bits [5:3] charge pump curre nt. varies the current that drives the low-pass filter. (see pll description.) 04h r/w 7:3 10000*** phase adjust adc cloc k phase adjustment. larger values mean more delay. (1 lsb = t/32) 05h r/w 7:0 10000000 clamp placement places the clamp signal an integer number of clock periods after the trailing edge of the hsync signal. 06h r/w 7:0 10000000 clamp duration number of clock periods that the clamp signal is actively clamping. 07h r/w 7:0 00100000 hsync output pulsewidth sets the number of pixel clocks that hsout will remain active. 08h r/w 7:0 10000000 red gain 09h r/w 7:0 10000000 green gain 0ah r/w 7:0 10000000 blue gain controls adc input range (contrast) of each respective channel. greater values give less contrast. 0bh r/w 7:1 1000000* red offset 0ch r/w 7:1 1000000* green offset 0dh r/w 7:1 1000000* blue offset controls dc offset (brightness) of each respective channel. greater values decrease brightness. 0eh r/w 7:0 0******* sync control bit 7 C hsync polarity override. (logic 0 = polarity determined by chip, logic 1 = polarity set by bit 6 in register 0eh.) *1****** bit 6 C hsync input polarity. indicate s polarity of incoming hsync signal to the pll. (logic 0 = active low, logic 1 = active high.) **0***** bit 5 C hsync output polarity. (logic 0 = logic high sync, logic 1 = logic low sync.) ***0**** bit 4 C active hsync override. if set to logic 1, the user can select the hsync to be used via bit 3. if set to logic 0, the active interface is selected via bit 6 in register 14h. ****0*** bit 3 C active hsync select. logic 0 selects hsync as the active sync. logic 1 selects sync-on-green as the active sync. note that the indicated hsync will be used only if bit 4 is set to logic 1 or if both syncs are active. (bits 1, 7 = logic 1 in register 14h.) *****0** bit 2 C vsync output invert. (logic 1 = no invert, logic 0 = invert.) ******0* bit 1 C active vsync override. if set to logic 1, the user can select the vsync to be used via bit 0. if set to logic 0, the active interface is selected via bit 3 in register 14h. *******0 bit 0 C active vsync select. logic 0 selects raw vsync as the output vsync. logic 1 selects sync separate d vsync as the o utput vsync. note that the indicated vsync will be used only if bit 1 is set to logic 1. 0fh r/w 7:1 0******* bit 7 C clamp function. chooses between hsync for clamp signal or another external signal to be used for clamping. (logic 0 = hsync, logic 1 = clamp.) *1****** bit 6 C clamp polarity. valid only with external clamp signal. (logic 0 = active high, logic 1 selects active low.) **0***** bit 5 C coast select. logic 0 selects th e coast input pins to be used for the pll coast. logic 1 selects vsyn c to be used for the pll coast. ***0**** bit 4 C coast polarity override. (logic 0 = polarity determined by chip, logic 1 = polarity set by bit 3 in register 0fh.) ****1*** bit 3 C coast polarity. selects polarity of external coast signal. (logic 0 = active low, logic 1 = active high.) *****1** bit 2 C seek mode override. (logic 1 = allow low power mode, logic 0 = disallow low power mode.) ******1* bit 1 C pwrdn. full chip power-down, active low. (logic 0 = full chip power-down, logic 1 = normal.) 10h r/w 7:3 10111*** sync-on-green threshold sync-on-green threshold. sets the voltage level of the sync-on-green slicers comparator.
ad9985 rev. 0 | page 18 of 32 hex address write and read or read only bits default value register name function *****0** bit 2 C red clamp select. logic 0 selects clamp to ground. logic 1 selects clamp to midscale (voltage at pin 37). ******0* bit 1 C green clamp select. logic 0 selects clamp to ground. logic 1 selects clamp to midscale (voltage at pin 37). *******0 bit 0 C blue clamp select. logic 0 selects clamp to ground. logic 1 selects clamp to midscale (voltage at pin 37). 11h r/w 7:0 00100000 sync separator threshold sync separator threshold. sets how many internal 5 mhz clock periods the sync separator will count to before toggling high or low. this should be set to some number gr eater than the maximum hsync or equalization pulsewidth. 12h r/w 7:0 00000000 pre-coast pre-coast. sets the number of hsync periods that coast becomes active prior to vsync. 13h r/w 7:0 00000000 post-coast post-coast. sets the number of hs ync periods that coast stays active following vsync. 14h ro 7:0 sync detect bit 7 C hsync detect. it is set to logic 1 if hsync is present on the analog interface; otherwise it is set to logic 0. bit 6 C ahs: active hsync. this bit indicates which analog hsync is being used. (logic 0 = hsync input pin, logic 1 = hsync from sync-on- green.) bit 5 C input hsync polarity detect. (logic 0 = active low, logic 1 = active high.) bit 4 C vsync detect. it is set to logi c 1 if vsync is present on the analog interface; otherwise it is set to logic 0. bit 3 C avs: active vsync. this bi t indicates which analog vsync is being used. (logic 0 = vsync input pin, logic 1 = vsync from sync separator.) bit 2 C output vsync polarity detect . (logic 0 = active low, logic 1 = active high.) bit 1 C sync-on-green detect. it is se t to logic 1 if sync is present on the green video input; otherwise it is set to 0. bit 0 C input coast polarity detect. (logic 0 = active low, logic 1 = active high.) 15h r/w 7:2 111111** reserved bits [7:2] reserved for future use. must be written to 111111 for proper operation. 1 ******1* output formats bit 1 C 4:2:2 output formatting mode (logic 0 = 4:2:2 mode, logic 1= 4:4:4 mode) 0 *******1 reserved bit 0 C must be set to 0 for proper operation. 16h r/w 7:0 test register reserved for future use. 17h ro 7:0 test register reserved for future use. 18h ro 7:0 test register reserved for future use. 19h r/w 7:0 00000100 red target code target co de for auto offset operation. 1ah r/w 7:0 00000100 green target code target code for auto offset operation. 1bh r/w 7:0 00000100 blue target code target code for auto offset operation. 1ch r/w 7:0 00010001 reserved must be written to 11h for proper operation. 1dh r/w 7 0******* auto offset enable enables the auto offset circuitry. 6 *0****** hold auto offset holds the offset o utput of the auto offset at the current value. 5:2 **1001** reserved must be written to 9 for proper operation. 1:0 ******10 update mode changes the update rate of the auto offset. 1eh r/w 7:0 0000**** test register must be set to default value. *the ad9985 updates the pll divide ratio only wh en the lsbs are written to (register 02h).
ad9985 rev. 0 | page 19 of 32 2-wire serial control register detail chip identification 00 7C0 chip revision an 8-bit register that represents the silicon revision. pll divider control 01 7C0 pll divide ratio msbs the 8 most significant bits of the 12-bit pll divide ratio plldiv. the operational divide ratio is plldiv + 1. the pll derives a master clock from an incoming hsync signal. the master clock frequency is then divided by an integer value, such that the output is phase-locked to hsync. this plldiv value determines the number of pixel times (pixels plus horizontal blanking overhead) per line. this is typically 20% to 30% more than the number of active pixels in the display. the 12-bit value of the pll divider supports divide ratios from 2 to 4095. the higher the value loaded in this register, the higher the resulting clock frequency with respect to a fixed hsync frequency. vesa has established some standard timing specifications that assist in determining the value for plldiv as a function of horizontal and vertical display resolution and frame rate (table 9). however, many computer systems do not conform precisely to the recommendations, and these numbers should be used only as a guide. the display system manufacturer should provide automatic or manual means for optimizing plldiv. an incorrectly set plldiv will usually produce one or more vertical noise bars on the display. the greater the error, the greater the number of bars produced. the power-up default value of plldiv is 1693 (plldivm = 69h, plldivl = dxh). the ad9985 updates the full divide ratio only when the lsbs are changed. writing to the msb by itself will not trigger an update. 02 7C4 pll divide ratio lsbs the 4 least significant bits of the 12-bit pll divide ratio plldiv. the operational divide ratio is plldiv + 1. the power-up default value of plldiv is 1693 (plldivm = 69h, plldivl = dxh). the ad9985 updates the full divide ratio only when this register is written to. clock generator control 03 7C6 vco range select two bits that establish the operating range of the clock generator. vcornge must be set to correspond with the desired operating frequency (incoming pixel rate). the pll gives the best jitter performance at high frequencies. for this reason, to output low pixel rates and still get good jitter performance, the pll actually operates at a higher frequency but then divides down the clock rate afterwards. table 11 shows the pixel rates for each vco range setting. the pll output divisor is automatically selected with the vco range setting. table 11. vco ranges pixel clock range (mhz) pv1 pv0 ad9985kstz ad9985bstz 0 0 12C32 12C30 0 1 32C64 30C60 1 0 64C110 60C110 1 1 110C140 the power-up default value is 01. 03 5C3 current charge pump current three bits that establish the current driving the loop filter in the clock generator. table 12. charge pump currents current current (a) 000 50 001 100 010 150 011 250 100 350 101 500 110 750 111 1500 current must be set to correspond with the desired operating frequency (incoming pixel rate). the power-up default value is current = 001. 04 7C3 clock phase adjust a 5-bit value that adjusts the sampling phase in 32 steps across one pixel time. each step represents an 11.25 shift in sampling phase. the power-up default value is 16.
ad9985 rev. 0 | page 20 of 32 clamp timing 05 7C0 clamp placement an 8-bit register that sets the position of the internally generated clamp. when clamp function (register 0fh, bit 7) = 0, a clamp signal is generated internally, at a position established by the clamp placement and for a duration set by the clamp duration. clamping is started (clamp placement) pixel periods after the trailing edge of hsync. the clamp placement may be programmed to any value between 1 and 255. the clamp should be placed during a time that the input signal presents a stable black-level reference, usually the back porch period between hsync and the image. when clamp function = 1, this register is ignored. 06 7C0 clamp duration an 8-bit register that sets the duration of the internally generated clamp. for the best results, the clamp duration should be set to include the majority of the black reference signal time that follows the hsync signal trailing edge. insufficient clamping time can produce brightness changes at the top of the screen, and a slow recovery from large changes in the average picture level (apl), or brightness. when clamp function = 1, this register is ignored. hsync pulsewidth 07 7C0 hsync output pulsewidth an 8-bit register that sets the duration of the hsync output pulse. the leading edge of the hsync output is triggered by the internally generated, phase-adjusted pll feedback clock. the ad9985 then counts a number of pixel clocks equal to the value in this register. this triggers the trailing edge of the hsync output, which is also phase adjusted. input gain 08 7C0 red channel gain adjust an 8-bit word that sets the gain of the red channel. the ad9985 can accommodate input signals with a full-scale range of between 0.5 v and 1.0 v p-p. setting redgain to 255 corresponds to a 1.0 v input range. a redgain of 0 establishes a 0.5 v input range. note that increasing redgain results in the picture having less contrast (the input signal uses fewer of the available converter codes). see figure 4. 09 7C0 green channel gain adjust an 8-bit word that sets the gain of the green channel. see redgain (08). 0a 7C0 blue channel gain adjust an 8-bit word that sets the gain of the blue channel. see redgain (08). input offset 0b 7C1 red channel offset adjust this and the following two offset registers have two modes of operation. one mode is when the auto offset function is turned off (manual mode) and the other is when auto offset is turned on. when in manual offset adjustment mode (auto offset turned off) this register behaves exactly like the ad9883a. it is a 7-bit offset binary word that sets the dc offset of the red channel. one lsb of offset adjustment equals approximately one lsb change in the adc offset. therefore, the absolute magnitude of the offset adjustment scales as the gain of the channel is changed. a nominal setting of 63 results in the channel nominally clamping the back porch (during the clamping interval) to code 00. an offset setting of 127 results in the channel clamping to code 64 of the adc. an offset setting of 0 clamps to code C63 (off the bottom of the range). increasing the value of red offset decreases the brightness of the channel. when in auto offset mode, the value in this register is digitally added to the red channel adc output. the purpose of doing this is to match a benefit that is present with manual offset adjustment. adjusting these registers is an easy way to make brightness adjustments. although some signal range is lost with this method, it has proven to be a very popular function. in order to be able to increase and decrease brightness, the values in these registers in this mode are signed twos complement (as opposed to manual mode where the values in this register are binary). the digital adder is used only when in auto offset mode. although it cannot be disabled, setting this register to all 0s will effectively disable it by always adding 0. 0c 7C1 green channel offset adjust this register works exactly like the red channel offset adjust register (0bh), except it is for the green channel. 0d 7C1 blue channe l offset adjust this register works exactly like the red channel offset adjust register (0bh), except it is for the blue channel.
ad9985 rev. 0 | page 21 of 32 mode control 1 0e 7 hsync input polarity override this register is used to override the internal circuitry that determines the polarity of the hsync signal going into the pll. table 13. hsync input polarity override settings override bit function 0 hsync polarity determined by chip 1 hsync polarity determined by user the default for hsync polarity override is 0 (polarity determined by chip). 0e 6 hspol hsync input polarity a bit that must be set to indicate the polarity of the hsync signal that is applied to the pll hsync input. table 14. hsync input polarity settings hspol function 0 active low 1 active high active low means the leading edge of the hsync pulse is negative going. all timing is based on the leading edge of hsync, which is the falling edge. the rising edge has no effect. active high is inverted from the traditional hsync, with a positive-going pulse. this means that timing will be based on the leading edge of hsync, which is now the rising edge. the device will operate if this bit is set incorrectly, but the internally generated clamp position, as established by clamp placement (register 05h), will not be placed as expected, which may generate clamping errors. the power-up default value is hspol = 1. 0e 5 hsync output polarity this bit determines the polarity of the hsync output and the sog output. table 15 shows the effect of this option. sync indicates the logic state of the sync pulse. table 15. hsync output polarity settings setting sync 0 logic 1 (positive polarity) 1 logic 0 (negative polarity) the default setting for this register is 0. 0e 4 active hsync override this bit is used to override the automatic hsync selection, to override, set this bit to logic 1. when overriding, the active hsync is set via bit 3 in this register. table 16. active hsync override settings override result 0 autodetermines the active interface 1 override, bit 3 determines the active interface the default for this register is 0. 0e 3 active hsync select this bit is used under two conditions. it is used to select the active hsync when the override bit is set (bit 4). alternately, it is used to determine the active hsync when not overriding but both hsyncs are detected. table 17. active hsync select settings select result 0 hsync input 1 sync-on-green input the default for this register is 0. 0e 2 vsync output invert this bit inverts the polarity of the vsync output. table 18 shows the effect of this option. table 18. vsync output invert settings setting vsync output 0 invert 1 no invert the default setting for this register is 0. 0e 1 active vsync override this bit is used to override the automatic vsync selection. to override, set this bit to logic 1. when overriding, the active interface is set via bit 0 in this register. table 19. active vsync override settings override result 0 autodetermines the active vsync 1 override, bit 0 determines the active vsync the default for this register is 0. 0e 0 active vsync select this bit is used to select the active vsync when the override bit is set (bit 1). table 20. active vsync select settings select result 0 vsync input 1 sync separator output the default for this register is 0.
ad9985 rev. 0 | page 22 of 32 0f 7 clamp input signal source this bit determines the source of clamp timing. table 21. clamp input signal source settings clamp function function 0 internally generated clamp signal 1 externally provided clamp signal a 0 enables the clamp timing circuitry controlled by clamp placement and clamp duration. the clamp position and duration is counted from the leading edge of hsync. a 1 enables the external clamp input pin. the three channels are clamped when the clamp signal is active. the polarity of clamp is determined by the clamp polarity bit (register 0fh, bit 6). the power-up default value is clamp function = 0. 0f 6 clamp input signal polarity this bit determines the polarity of the externally provided clamp signal. table 22. clamp input signal polarity settings clamp function function 1 active low 0 active high logic 1 means that the circuit will clamp when clamp is low, and it will pass the signal to the adc when clamp is high. logic 0 means that the circuit will clamp when clamp is high, and it will pass the signal to the adc when clamp is low. the power-up default value is clamp polarity = 1. 0f 5 coast select this bit is used to select the active coast source. the choices are the coast input pin or vsync. if vsync is selected, the additional decision of using the vsync input pin or the output from the sync separator needs to be made (register 0e, bits 1, 0). table 23. power-down settings select result 0 coast input pin 1 vsync (see above text) 0f 4 coast input polarity override this register is used to override the internal circuitry that determines the polarity of the coast signal going into the pll. table 24. coast input polarity override settings override bit result 0 determined by chip 1 determined by user the default for coast polarity override is 0. 0f 3 coast input polarity this bit indicates the polarity of the coast signal that is applied to the pll coast input. table 25. coast input polarity settings coast polarity function 0 active low 1 active high active low means that the clock generator will ignore hsync inputs when coast is low, and continue operating at the same nominal frequency until coast goes high. active high means that the clock generator will ignore hsync inputs when coast is high, and continue operating at the same nominal frequency until coast goes low. this function needs to be used along with the coast polarity override bit (bit 4). the power-up default value is 1. 0f 2 seek mode override this bit is used to either allow or disallow the low power mode. the low power mode (seek mode) occurs when there are no signals on any of the sync inputs. table 26. seek mode override settings select result 1 allow seek mode 0 disallow seek mode the default for this register is 1. 0f 1 pwrdn this bit is used to put the chip in full power-down. see the power management section for details of which blocks are powered down. table 27. power-down settings select result 0 power-down 1 normal operation 10 7-3 sync-on-green slicer threshold this register allows the comparator threshold of the sync-on-green slicer to be adjusted. this register adjusts it in steps of 10 mv, with the minimum setting equaling 10 mv (11111) and the maximum setting equaling 330 mv (00000). the default setting is 23, which corresponds to a threshold value of 100 mv; for a threshold of 150 mv, the setting should be 18.
ad9985 rev. 0 | page 23 of 32 10 2 red clamp select this bit determines whether the red channel is clamped to ground or to midscale. for rgb video, all three channels are referenced to ground. for ycbcr (or yuv), the y channel is referenced to ground, but the cbcr channels are referenced to midscale. clamping to midscale actually clamps to pin 37. table 28. red clamp select settings clamp function 0 clamp to ground 1 clamp to midscale (pin 37) the default setting for this register is 0. 10 1 green clamp select this bit determines whether the green channel is clamped to ground or to midscale. table 29. green clamp select settings clamp function 0 clamp to ground 1 clamp to midscale (pin 37) the default setting for this register is 0. 10 0 blue clamp select this bit determines whether the blue channel is clamped to ground or to midscale. table 30. blue clamp select settings clamp function 0 clamp to ground 1 clamp to midscale (pin 37) the default for this register is 0. 11 7C0 sync separator threshold this register is used to set the responsiveness of the sync separator. it sets how many internal 5 mhz clock periods the sync separator must count to before toggling high or low. it works like a low-pass filter to ignore hsync pulses in order to extract the vsync signal. this register should be set to some number greater than the maximum hsync pulsewidth. note that the sync separator threshold uses an internal dedicated clock with a frequency of approximately 5 mhz. the default for this register is 32. 12 7C0 pre-coast this register allows the coast signal to be applied prior to the vsync signal. this is necessary in cases where pre-equalization pulses are present. the step size for this control is one hsync period. the default is 0. 13 7C0 post-coast this register allows the coast signal to be applied following the vsync signal. this is necessary in cases where post-equalization pulses are present. the step size for this control is one hsync period. the default is 0. 14 7 hsync detect this bit is used to indicate when activity is detected on the hsync input pin (pin 30). if hsync is held high or low, activity will not be detected. table 31. hsync detection results detect function 0 no activity detected 1 activity detected the sync processing block diagram (figure 14) shows where this function is implemented. 14 6 ahs C active hsync this bit indicates which hsync input source is being used by the pll (hsync input or sync-on-green). bits 7 and 1 in this register determine which source is used. if both hsync and sog are detected, the user can determine which has priority via bit 3 in register 0eh. the user can override this function via bit 4 in register 0eh. if the override bit is set to logic 1, this bit will be forced to whatever the state of bit 3 in register 0eh is set to. table 32. active hsync results bit 7 bit 1 bit 4, (hsync (sog reg 0eh detect) detect) (override) ahs 0 0 0 bit 3 in 0eh 0 1 0 1 1 0 0 0 1 1 0 bit 3 in 0eh x x 1 bit 3 in 0eh ahs = 0 means use the hsync pin input for hsync. ahs = 1 means use the sog pin input for hsync. the override bit is in register 0eh, bit 4. 14 5 detected hsync input polarity status this bit reports the status of the hsync input polarity detection circuit. it can be used to determine the polarity of the hsync input. the detection circuits location is shown in the sync processing block diagram (figure 14).
ad9985 rev. 0 | page 24 of 32 table 33. detected hsync input polarity status hsync polarity status result 0 negative 1 positive 14 4 vsync detect this bit is used to indicate when activity is detected on the vsync input pin (pin 31). if vsync is held steady high or low, activity will not be detected. table 34. vsync detection results detect function 0 no activity detected 1 activity detected the sync processing block diagram (figure 14) shows where this function is implemented. 14 3 avs C active vsync this bit indicates which vsync source is being used: the vsync input or output from the sync separator. bit 4 in this register determines which is active. if both vsync and sog are detected, the user can determine which has priority via bit 0 in register 0eh. the user can override this function via bit 1 in register 0eh. if the override bit is set to logic 1, this bit will be forced to whatever the state of bit 0 in register 0eh is set to. table 35. active vsync results bit 4, reg 14h bit 1, reg 0eh (vsync detect) (override) avs 1 0 0 0 0 1 x 1 bit 0 in 0eh avs = 0 me ans vsy nc input. avs = 1 me ans sy nc s ep arator. the override bit is in register 0eh, bit 1. 14 2 detected vsync output polarity status this bit reports the status of the vsync output polarity detection circuit. it can be used to determine the polarity of the vsync output. the detection circuits location is shown in the sync processing block diagram (figure 14). table 36. detected vsync output polarity status vsync polarity status result 0 active low 1 active high 14 1 sync-on-green detect this bit is used to indicate when sync activity is detected on the sync-on-green input pin (pin 49). table 37. sync-on-green detection results detect function 0 no activity detected 1 activity detected the sync processing block diagram (figure 14) shows where this function is implemented. 14 0 detected coast polarity status this bit reports the status of the coast input polarity detection circuit. it can be used to determine the polarity of the coast input. the detection circuits location is shown in the sync processing block diagram (figure 14). table 38. detected coast input polarity status polarity status result 0 coast polarity negative 1 coast polarity positive this indicates that bit 1 of register 5 is the 4:2:2 output mode select bit. 15 1 4:2:2 output mode select this bit configures the output data in 4:2:2 mode. this mode can be used to reduce the number of data lines used from 24 down to 16 for applications using yuv, ycbcr, or ypbpr graphics signals. a timing diagram for this mode is shown in figure 11. recommended input and output configurations are shown in table 39. table 39. 4:2:2 output mode select select output mode 0 4:2:2 1 4:4:4 table 40. 4:2:2 input/output configuration input channel connection output format red v u/v green y y blue u high impedance 19 7:0 red target code this specifies the targeted value of the final offset for the red channel when auto offset is employed (register 0x1d bit 7 = 1). default is 4. 1a 7:0 green target code this specifies the targeted value of the final offset for the green channel when auto offset is employed (register 0x1d bit 7 = 1). default is 4.
ad9985 rev. 0 | page 25 of 32 1b 7:0 blue target code this specifies the targeted value of the final offset for the blue channel when auto offset is employed (register 0x1d bit 7 = 1). default is 4. 1d 7 auto offset enable enables the auto offset circuitry. default is 0. 1d 6 hold auto offset holds the offset output of the auto offset at the current value. default is 0. 1d 1:0 update mode changes the update rate of the auto offset. default is 10. table 41. auto offset update rate update mode auto-offset update timing 00 every clamp cycle. 01 every 16 clamp cycles. 10 every 64 clamp cycles.
ad9985 rev. 0 | page 26 of 32 2-wire serial control port a 2-wire serial control interface (i 2 c) is provided. up to two ad9985 devices may be connected to the 2-wire serial interface, with each device having a unique address. the 2-wire serial interface comprises a clock (scl) and a bidirectional data (sda) pin. the analog flat panel interface acts as a slave for receiving and transmitting data over the serial interface. when the serial interface is not active, the logic levels on scl and sda are pulled high by external pull-up resistors. data received or transmitted on the sda line must be stable for the duration of the positive-going scl pulse. data on sda must change only when scl is low. if sda changes state while scl is high, the serial interface interprets that action as a start or stop sequence. there are five components to serial bus operation: ? start signal ? slave address byte ? base register address byte ? data byte to read or write ? stop signal when the serial interface is inactive (scl and sda are high), communications are initiated by sending a start signal. the start signal is a high-to-low transition on sda while scl is high. this signal alerts all slaved devices that a data transfer sequence is coming. the first eight bits of data transferred after a start signal comprise a 7-bit slave address (the first seven bits) and a single r/ w bit (the eighth bit). the r/ w bit indicates the direction of data transfer, read from (1) or write to (0) the slave device. if the transmitted slave address matches the address of the device (set by the state of the sa1-0 input pins in table 42), the ad9985 acknowledges by bringing sda low on the ninth scl pulse. if the addresses do not match, the ad9985 does not acknowledge. table 42. serial port addresses bit 7 bit 6 bit 5 bit 4 bit 3 bit 2 bit 1 a 6 a 5 a 4 a 3 a 2 a 1 a 0 (msb) 1 0 0 1 1 0 0 1 0 0 1 1 0 1 data transfer via serial interface for each byte of data read or written, the msb is the first bit of the sequence. if the ad9985 does not acknowledge the master device during a write sequence, the sda remains high so the master can generate a stop signal. if the master device does not acknowl- edge the ad9985 during a read sequence, the ad9985 interprets this as end of data. the sda remains high so the master can generate a stop signal. writing data to specific control registers of the ad9985 requires that the 8-bit address of the control register of interest be written after the slave address has been established. this control register address is the base address for subsequent write opera- tions. the base address autoincrements by one for each byte of data written after the data byte intended for the base address. sda scl t buff t stah t dho t dsu t dal t dah t stasu t stosu 04799-0-012 figure 12. serial port read/write timing
ad9985 rev. 0 | page 27 of 32 data is read from the control registers of the ad9985 in a similar manner. reading requires two data transfer operations: the base address must be written with the r/w bit of the slave address byte low to set up a sequential read operation. reading (the r/ w bit of the slave address byte high) begins at the previously established base address. the address of the read register autoincrements after each byte is transferred. to terminate a read/write sequence to the ad9985, a stop signal must be sent. a stop signal comprises a low-to-high transition of sda while scl is high. a repeated start signal occurs when the master device driving the serial interface generates a start signal without first generating a stop signal to terminate the current communi- cation. this is used to change the mode of communication (read, write) between the slave and master without releasing the serial interface lines. serial interface read/write examples write to one control register ? start signal ? slave address byte (r/ w bit = low) ? base address byte ? data byte to base address ? stop signal write to four consecutive control registers ? start signal ? slave address byte (r/ w bit = low) ? base address byte ? data byte to base address ? data byte to (base address + 1) ? data byte to (base address + 2) ? data byte to (base address + 3) ? stop signal read from one control register ? start signal ? slave address byte (r/ w bit = low) ? base address byte ? start signal ? slave address byte (r/ w bit = high) ? data byte from base address ? stop signal read from four consecutive control registers ? start signal ? slave address byte (r/ w bit = low) ? base address byte ? start signal ? slave address byte (r/ w bit = high) ? data byte from base address ? data byte from (base address + 1) ? data byte from (base address + 2) ? data byte from (base address + 3) ? stop signal bit 7 ack bit 6 bit 5 bit 4 bit 3 bit 2 bit 1 bit 0 sda s cl 04799-0-014 figure 13. serial interfacetypical byte transfer
ad9985 rev. 0 | page 28 of 32 sync stripper activity detect negative peak clamp comp sync sog hsync in activity detect mux 2 hsync out pixel clock mux 1 sync separator integrator vsync sog out hsync out vsync out mux 4 vsync in 1/s pll hsync activity detect ad9985 clock generator polarity detect polarity detect polarity detect mux 3 coast coast 04799-0-0015 figure 14. sync processing block diagram table 43. control of the sync block muxes via the serial register serial bus control mux no. control bit bit state result 1 and 2 0eh: bit 3 0 pass hsync 1 pass sync-on-green 3 0fh: bit 5 0 pass coast 1 pass vsync 4 0eh: bit 0 0 pass vsync 1 pass sync separator signal sync slicer the purpose of the sync slicer is to extract the sync signal from the green graphics channel. a sync signal is not present on all graphics systems, only those with sync-on-green. the sync signal is extracted from the green channel in a two-step process. first, the sog input is clamped to its negative peak (typically 0.3 v below the black level). next, the signal goes to a comparator with a variable trigger level, nominally 0.15 v above the clamped level. the sliced sync is typically a composite sync signal containing both hsync and vsync. sync separator a sync separator extracts the vsync signal from a composite sync signal. it does this through a low-pass filter-like or integrator-like operation. it works on the idea that the vsync signal stays active for a much longer time than the hsync signal, so it rejects any signal shorter than a threshold value, which is somewhere between an hsync pulsewidth and a vsync pulsewidth. the sync separator on the ad9985 is simply an 8-bit digital counter with a 5 mhz clock. it works independently of the polarity of the composite sync signal. (polarities are determined elsewhere on the chip.) the basic idea is that the counter counts up when hsync pulses are present. but since hsync pulses are relatively short in width, the counter only reaches a value of n before the pulse ends. it then starts counting down, eventually reaching 0 before the next hsync pulse arrives. the specific value of n will vary for different video modes, but will always be less than 255. for example, with a 1 s width hsync, the counter will only reach 5 (1 s/200 ns = 5). when vsync is present on the composite sync, the counter will also count up. however, since the vsync signal is much longer, it will count to a higher number m. for most video modes, m will be at least 255. so, vsync can be detected on the composite sync signal by detecting when the counter counts to higher than n. the specific count that triggers detection (t) can be programmed through the serial register (11h). once vsync has been detected, there is a similar process to detect when it goes inactive. at detection, the counter first resets to 0, then starts counting up when vsync goes away. similar to the previous case, it will detect the absence of vsync when the counter reaches the threshold count (t). in this way, it will reject noise and/or serration pulses. once vsync is detected to be absent, the counter resets to 0 and begins the cycle again.
ad9985 rev. 0 | page 29 of 32 pcb layout recommendations the ad9985 is a high precision, high speed analog device. as such, to get the maximum performance from the part, it is important to have a well laid out board. the following is a guide for designing a board using the ad9985. analog interface inputs using the following layout techniques on the graphics inputs is extremely important. minimize the trace length running into the graphics inputs. this is accomplished by placing the ad9985 as close as possible to the graphics vga connector. long input trace lengths are undesirable because they pick up more noise from the board and other external sources. place the 75 ? termination resistors (see figure 3) as close to the ad9985 chip as possible. any additional trace length between the termination resistors and the input of the ad9985 increases the magnitude of reflections, which will corrupt the graphics signal. use 75 ? matched impedance traces. trace impedances other than 75 ? will also increase the chance of reflections. the ad9985 has very high input bandwidth (500 mhz). while this is desirable for acquiring a high resolution pc graphics signal with fast edges, it means that it will also capture any high frequency noise present. therefore, it is important to reduce the amount of noise that gets coupled to the inputs. avoid running any digital traces near the analog inputs. due to the high bandwidth of the ad9985, low-pass filtering the analog inputs can sometimes help to reduce noise. (for many applications, filtering is unnecessary.) experiments have shown that placing a series ferrite bead prior to the 75 ? termination resistor is helpful in filtering out excess noise. specifically, the part used was the #2508051217z0 from fair- rite, but each application may work best with a different bead value. alternately, placing a 100 ? to 120 ? resistor between the 75 ? termination resistor and the input coupling capacitor can also be beneficial. power supply bypassing it is recommended to bypass each power supply pin with a 0.1 f capacitor. the exception is when two or more supply pins are adjacent to each other. for these groupings of powers/ grounds, it is necessary to have only one bypass capacitor. the fundamental idea is to have a bypass capacitor within about 0.5 cm of each power pin. also, avoid placing the capacitor on the opposite side of the pc board from the ad9985, as that interposes resistive vias in the path. the bypass capacitors should be physically located between the power plane and the power pin. current should flow from the power plane to the capacitor to the power pin. do not make the power connection between the capacitor and the power pin. placing a via underneath the capacitor pads, down to the power plane, is generally the best approach. it is particularly important to maintain low noise and good stability of pv d (the clock generator supply). abrupt changes in pv d can result in similarly abrupt changes in sampling clock phase and frequency. this can be avoided by careful attention to regulation, filtering, and bypassing. it is highly desirable to provide separate regulated supplies for each of the analog circuitry groups (v d and pv d ). some graphic controllers use substantially different levels of power when active (during active picture time) and when idle (during horizontal and vertical sync periods). this can result in a measurable change in the voltage supplied to the analog supply regulator, which can in turn produce changes in the regulated analog supply voltage. this can be mitigated by regulating the analog supply, or at least pv d , from a different, cleaner power source (for example, from a 12 v supply). it is also recommended to use a single ground plane for the entire board. experience has repeatedly shown that the noise performance is the same or better with a single ground plane. using multiple ground planes can be detrimental because each separate ground plane is smaller, and long ground loops can result. in some cases, using separate ground planes is unavoidable. for those cases, it is recommended to at least place a single ground plane under the ad9985. the location of the split should be at the receiver of the digital outputs. for this case it is even more important to place components wisely because the current loops will be much longer (current takes the path of least resistance). an example of a current loop is shown in figure 15. a n a l o g g r o u n d p l a n e p o w e r p l a n e a d 9 8 8 3 a d i g i t a l o u t p u t t r a c e d i g i t a l g r o u n d p l a n e d i g i t a l d a t a r e c e i v e r 04799-0-016 figure 15. current loop
ad9985 rev. 0 | page 30 of 32 pll place the pll loop filter components as close to the filt pin as possible. do not place any digital or other high frequency traces near these components. use the values suggested in the data sheet with 10% tolerances or less. outputs (both data and clocks) try to minimize the trace length that the digital outputs have to drive. longer traces have higher capacitance, which requires more current, which causes more internal digital noise. shorter traces reduce the possibility of reflections. adding a series resistor of value 22 ? to 100 ? can suppress reflections, reduce emi, and reduce the current spikes inside of the ad9985. however, if 50 ? traces are used on the pcb, the data outputs should not need resistors. a 22 ? resistor on the datack output should provide good impedance matching that will reduce reflections. if series resistors are used, place them as close to the ad9985 pins as possible (although try not to add vias or extra length to the output trace in order to get the resistors closer). if possible, limit the capacitance that each of the digital outputs drives to less than 10 pf. this can easily be accomplished by keeping traces short and by connecting the outputs to only one device. loading the outputs with excessive capacitance will increase the current transients inside of the ad9985, creating more digital noise on its power supplies. digital inputs the digital inputs on the ad9985 were designed to work with 3.3 v signals, but are tolerant of 5.0 v signals. therefore, no extra components need to be added if using 5.0 v logic. any noise that gets onto the hsync input trace will add jitter to the system. therefore, minimize the trace length and do not run any digital or other high frequency traces near it. voltage reference bypass with a 0.1 f capacitor. place as close to the ad9985 pin as possible. make the ground connection as short as possible.
ad9985 rev. 0 | page 31 of 32 outline dimensions 1.45 1.40 1.35 0.15 0.05 61 60 1 80 20 41 21 40 top view (pins down) pin 1 seating plane view a 1.60 max 0.75 0.60 0.45 0.20 0.09 0.10 max coplanarity view a rotated 90 ccw seating plane 10 6 2 7 3.5 0 14.00 bsc sq 16.00 bsc sq 0.65 bsc 0.38 0.32 0.22 compliant to jedec standards ms-026-bec figure 16. 80-lead low profile quad flat package (lqfp) (st-80-2) dimensions shown in millimeters ordering guide model temperature range package description package ad9985kstz-110 1 0c to 70c lqfp st-80 ad9985kstz-140 1 0c to 70c lqfp st-80 ad9985bstz-110 1 C40c to +85c lqfp st-80 ad9985/pcb 25c evaluation board 1 z = pb-free part.
ad9985 rev. 0 | page 32 of 32 notes purchase of licensed i 2 c components of analog devices or one of its sublicensed associated companies conveys a license for the purchaser under the phi lips i 2 c patent rights to use these components in an i 2 c system, provided that the system conforms to the i 2 c standard specification as defined by philips. ? 2004 analog devices, inc. all rights reserved. trademarks and registered trademarks are the prop erty of their respective owners. d04799-0-5/04(0)

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